Solid-state regulated voltage supply

ABSTRACT

A solid-state (IC) regulated voltage supply compensated for effects of changes in temperature comprising first and second transistors operated at different current densities. Associated circuitry develops a voltage proportional to the ΔV BE  of the two transistors and having a positive temperature coefficient. This voltage is connected in series with the V BE  voltage of one of the two transistors, having a negative temperature coefficient, to produce a resultant voltage with nearly zero temperature coefficient. A feedback circuit responsive to current flow through the two transistors automatically adjusts the base voltages to maintain a predetermined ratio of current density for the two transistors. Other embodiments provide higher-level DC outputs and compensation for base current flow.

.Iadd.This is a continuation of U.S. Pat. application Ser. No. 799,760filed May 23, 1977, which is a Reissue application of U.S. Pat. No.3,887,638 dated June 3, 1975. .Iaddend.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to regulated DC voltage supplies. Moreparticularly, this invention relates to solid-state (IC) regulatorscapable of maintaining a substantially constant DC output voltage in theface of temperature variations.

2. Description of the Prior Art

Conventional prior-art regulated voltage supplies commonly have includedan internal reference source and an error amplifier arranged to comparethe reference voltage with a pre-set fraction of the regulated DC outputvoltage. The output of the error amplifier is directed to a controlelement, such as a controllable impedance or the like, arranged toadjust the output DC voltage so as to maintain the two compared voltagesequal. Fluctuations in the DC output voltage are thereby reduced.

In transistorized voltage-regulator circuits, the reference sourcetypically has been a Zener diode. However, as is known in the art, Zenerdiodes have certain inherent characteristics which undesirably restrictthe capability of a voltage regulator. An alternative type ofsolid-state regulator has been developed which does not use a Zenerdiode reference, relying instead on certain temperature-dependentcharacteristics of the base-to-emitter voltage (V_(BE)) of a transistor.

U.S. Pat. No. 3,617,859 discloses a circuit of the latter type whichincludes a diode-connected transistor operated at one current density,and a second transistor operated at a different current density. Thesetwo transistors are interconnected with associated circuitry so as todevelop a voltage proportional to the difference in the respectivebase-to-emitter voltages (ΔV_(BE)). This difference voltage has apositive temperature-coefficient (TC), and is connected in series withthe V_(BE) voltage of a third transistor, having a negative TC, toproduce a composite resultant voltage which serves as the output of theregulator. Since the temperature coefficients of the two individualvoltages are of opposite sign, the output voltage can be made relativelyinsensitive to temperature variations by proper choice of certainparameters.

Although such regulators based on the V_(BE) characteristic oftransistors have significant advantages, the circuit arrangementsproposed and used heretofore suffer from serious limitations. It is aprincipal object of the present invention to provide a solid-statevoltage regulator which avoids or significantly minimizes suchlimitations of prior art regulators.

SUMMARY OF THE INVENTION

In an exemplary embodiment of the present invention, to be described indetail hereinbelow, there is provided a two-transistor voltage-regulatorcircuit wherein the ratio of current densities of the two transistors isautomatically controlled to a predetermined value (different from unity)by a negative feedback arrangement. A voltage corresponding to theΔV_(BE) of the two transistors is developed, having a positive TC, andthis voltage is connected in series with the V_(BE) voltage of one ofthe two transistors, having a negative TC. The circuit parameters areselected so that the resultant combined voltage has a very lowtemperature coefficient. The regulator of this invention providesimportant advantages over previous regulators, as will be outlinedhereinbelow in describing specific embodiments of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram of one embodiment of the present invention;

FIG. 2 is a circuit diagram of a modified arrangement to provide higherregulated output voltages;

FIG. 3 shows a further circuit arrangement modified to reduce basecurrent effects; and

FIG. 4 is a circuit diagram of a voltage reference source includingmeans to establish bias levels and to provide current limiting.

DESCRIPTION OF PREFERRED EMBODIMENTS

Referring now to FIG. 1, there is shown a circuit diagram representingbasic components and interconnections of a regulated voltage source inaccordance with the present invention. The circuit includes a pair oftransistors Q₁ and Q₂ which are supplied with operating voltages bypositive and negative voltage lines 10 and 12. The emitter of theleft-hand transistor Q₂ is coupled through two series-connectedresistors R₂ and R₁ to the negative voltage line 12, and the emitter ofthe other transistor Q₁ is connected to the common junction 18 betweenthe two resistors.

The invention proceeds on the concept of (1) developing a first voltage,having a positive temperature coefficient (TC), (2) combining thatvoltage serially with a second voltage having a negative TC, and (3)relating the two temperature coefficients in a complementary sense suchthat the resultant composite voltage has a very low TC, approximatelyzero. To develop the positive TC voltage, the two transistors Q₁ and Q₂are controllably operated at markedly different current densities (i.e.,referring to density of current flowing through the emitters), and avoltage is produced proportional to the difference in the two transistorbase-to-emitter voltages, referred to as ΔV_(BE).

In the specific embodiment disclosed herein, transistor Q₂ is operatedat a smaller current density than the other transistor Q₁. Suchdifference in current densities can be produced (1) by using identicaltransistors operating at unequal currents, (2) by using transistorshaving unequal emitter areas operating at equal currents, or (3) by somecombination of the latter two arrangements. Simply by way of example, inthe described embodiment the emitter areas of the transistors Q₁ and Q₂are specified as A and nA respectively, with n being greater than one,and the currents through the two transistors are equal.

The transistor currents are forced to be equal by a negative feedbackarrangement comprising current-sensing means in the form ofequal-resistance load resistors R_(L1) and R_(L2) in the collectorcircuits of transistors Q₁ and Q₂. These load resistors develop voltageswhich are proportional to the respective collector currents, and whichare directed by leads 24, 26 to the input terminals of a high-gainoperational amplifier 28. The output of this amplifier is connected to acommon base line of the two transistors Q₁ and Q₂, and also to an outputterminal 30 presenting the DC output voltage of the regulator. Theamplifier 28 drives the common base line until the voltage drops acrossthe load resistors R_(L1) and R_(L2) are equal, thereby forcing thetransistor collector currents to be equal. By using well matchedtransistors Q₁ and Q₂, the emitter currents also will be equal.

Since the bases of transistors Q₁ and Q₂ are connected together, thedifference voltage Δ_(VBE) will appear across emitter resistor R₂, andthe current through that resistor thus will be directly proportional toΔV_(BE). The series connected resistor R₁ also carries this emittercurrent, and additionally carries the emitter current of the secondtransistor Q₁. Since the latter emitter current is directly proportionalto the first emitter current (actually equal, in this particularexample), it will be evident that the total current through resistor R₁,and hence the voltage across that resistor, also will be directlyproportional to ΔV_(BE).

It has previously been established that, for two transistors operatingat different current densities, the difference in base-to-emittervoltage is given by:

    ΔV.sub.BE =kT/q ln J.sub.1 /J.sub.2

where T is absolute temperature, k is Boltzman's constant, q is thecharge of an electron, and J₁ /J₂ is the ratio of the transistor currentdensities. Accordingly, the voltage developed across resistor R₁ isindependent of absolute emitter current, and is a linear function ofabsolute temperature with a positive temperature coefficient.

As is evident from the circuit lay-out, the voltage across resistor R₁is in series with the V_(BE) voltage of transistor Q₁ and the resultantcomposite voltage constitutes the DC output voltage on terminal 30.Since V_(BE) has a negative temperature coefficient, changes in thatvoltage with temperature tend to complement the positive TC changes inthe voltage across resistor R₁.

To approach zero TC, the output voltage at the regulator terminal 30,with respect to the negative voltage line 12, should be setapproximately to the value of the energy band-gap voltage (V_(GO)),extrapolated to 0°. For silicon, this extrapolated voltage is 1.205V. Aslightly higher voltage produces superior results. It can be shownmathematically, based on certain reasonable assumptions, that for zeroTC the output voltage should be set at:

    V.sub.OUT =V.sub.GO +(m-1)kTo/q

where m≅1.5 and To is the nominal operating temperature.

This voltage V_(OUT) can be adjusted to the desired value by properselection of resistor R₁, such that the resistive voltage dropcomplements the V_(BE) of Q₁ to optimize the total (sum) voltage forzero TC.

When the DC output voltage (V_(OUT)) at terminal 30 drops below thepre-established optimal level, the ratio of collector currents I₂ /I₁ islarger than the ratio of load resistors R_(L1) /R_(L2) (i.e., largerthan one), so that the input to amplifier 28 is positive. This causesthe amplifier output to increase, so as to return the voltage V_(OUT)back up to the optimal level. If the DC output rises above optimal, thefeedback action of amplifier 28 will have the opposite effect. Thus thevoltage-control circuit continuously holds the DC output voltage at theproper level to provide a very low overall temperature coefficient,close to zero.

In some applications, DC output voltages higher than the energy band-gapvoltage may be required. FIG. 2 shows an arrangement for that purpose.The basic operation of this circuit is similar to that of FIG. 1, andlike reference numerals are used throughout for corresponding elements.However, FIG. 2 differs in that the output of amplifier 28 is connectedto a voltage-dividing network comprising two series-connected resistorshR₃ and R₃. The common junction terminal 32 of these resistors providesa voltage which is a predetermined fraction of V_(OUT), and this voltageis directed to the commonly connected bases of transistor Q₁ and Q₂.

As before, the amplifier 28 drives the transistor bases until theircollector currents are equal. By proper selection of circuit parameters,the reference voltage (V_(REF)) at this stable point can be set to beoptimum for achieving zero TC. The output voltage (V_(OUT)) then will besome predetermined multiple of V_(REF), specifically (h+1)·V_(REF).

This arrangement of FIG. 2 provides a quite accurate result. It isdegraded only a small amount due to the base current of the transistors.This base current is relatively low, and in any event the positive TC ofthe transistor beta tends to act with the positive TC of the emittercurrent to stabilize the base current and reduce any drift.

Where further reduction of such small drift effects may be desirable, acontrolled beta PNP may be used to reflect the base current from a pairmatched to Q₁ and Q₂ and connected in cascode with them into the base ofQ₁ and Q₂. Alternatively, the base of transistors Q₁ and Q₂ can beconnected together through a resistor R₄, as illustrated in FIG. 3.Here, the voltage across resistor R₂ is no longer ΔV_(BE), since thebases are no longer at the same voltage. However, it can be shown thatthis arrangement may, with matched betas, produce the basic regulationof the FIG. 2 embodiment, but with reduced drift due to base current,providing R₄ is selected such that:

    R.sub.4 =hR.sub.3 /h+1×R.sub.2 /R.sub.1

To take into account the possible effects of base spreading resistanceof the two transistors, further analysis indicates that thebase-connecting resistor R₄ should be selected such that:

    R.sub.4 =R.sub.b1 C.sub.1 (1 +1/C.sub.2 (C.sub.1 +1))-R.sub.b2

where R_(b1) and R_(b2) are the base spreading resistances oftransistors Q₁ and Q₂ ; C₁ =I_(e1) /I_(e2) (emitter currents of Q₁ andQ₂); and C₂ =R₁ /R₂

The above-derived expression for R₄ also indicates that the use of abase-connecting resistor may be helpful in the basic circuitconfiguration of FIG. 1. For R₄ to be zero, R_(b2) must be larger thanR_(b1) ; typically, however, Q₂ is the larger transistor with a very lowbase resistance, and design considerations thus suggest that the baseresistance of Q₁ should be minimized. It may be possible to correct forthe effect with a pinched base resistor in series with the largetransistor.

Voltage-regulated supplies in accordance with the present invention havea number of important and beneficial features. Foremost, such voltagesupplies provide a highly stable output voltage in the face of changingambient temperature. Only two matched active elements are required,rather than three as in the above-identified U.S. Pat. No. 3,617,859.Advantageously, the reference voltage in the disclosed circuits appearsin the control loop at a point with a high impedance, so that it canreadily be driven. Moreover, the reference voltage may be multiplied asdesired to produce output voltages higher than the band-gap voltage, bymeans of a single control loop, and without stacking junctions. In theFIG. 2 configuration, the reference voltage can first be adjusted tominimize temperature coefficient, and then the output voltage canseparately be adjusted to a predetermined voltage without affecting thetemperature coefficient. The basic circuit is convenient to trim byadjusting a single resistor (R₁). Finite beta and beta drift does notresult in uncorrectable errors; only beta matching is required.

Referring now to FIG. 4, there is shown a voltage reference sourceincluding transistors Q₁ and Q₂ used to establish the reference voltagein the manner generally as described hereinabove. In this circuit, thesetransistors are driven so that they will operate at equal collectorcurrents. Neglecting R23, for the moment, the bases of these transistorsare driven from the circuit output by the voltage divider consisting ofR31 and R24. The output current is provided by Darlington-connectedtransistors Q₄ and Q₇, which draw operating current from the inputvoltage terminal. The base of Q₄ is driven by a bias current from Q₁₈.

The circuit output voltage is controlled by adjusting the base voltageof Q₄, so that Q₄ and Q₇ form a voltage follower. A voltage dropprovided by Q₃ approximately matches the V_(BE) of Q₄, with R27 and Q₁₅providing a voltage drop matching other circuit voltages. The basevoltage of Q₄ is controlled by the emitter follower Q₁₂ which is drivenby Q₁ and Q₁₄.

In operation the collector current of Q₂ drives the base of Q₁₁negative. Acting as an emitter follower, Q₁₁ turns on Q₁₃ and drives ituntil its collector current approximately equals the collector currentof Q₂. The base of Q₁₃ connects to Q₁₄, a matching transistor. Since R25and R26 are also matched, the collector current of Q₁₄ willapproximately equal that of Q₁₃ and hence of Q₂. If the collectorcurrent of Q₂ exceeds the collector current of Q₁, Q₁₄ will drive thebase of Q₁₂ positive. Alternatively, if the collector current of Q₁exceeds the collector current of Q₂, it will also exceed that of Q₁₄ andwill, therefore, drive the base of Q₁₂ negative. The circuit outputvoltage will follow the base voltage at Q₁₂ as previously explained.

The emitter area of Q₂ is eight times larger than that of Q₁. When thevoltage at the base of Q₁ and Q₂ is low, the current through R21 and R22is low. The resulting voltage drop across R22 will be low, and thebase-emitter voltages of Q₁ and Q₂ will be nearly equal. As a result ofthe area ratio mismatch the emitter current in Q₂ will be nearly eighttimes the current in Q₁. This current mismatch will cause Q₁₄ to drivethe base of Q₁₂, and, thereby, the output - positive.

If the base voltage applied to Q₁ and Q₂ is made larger, the currentthrough R21 and R22 will also be larger. At a sufficiently high basevoltage the voltage drop across R22 will limit the current in Q₂, and itwill drop below the current in Q₁. The excess collector current in Q₁will drive the base of Q₁₂ negative, and with it the circuit output.

Between these two extremes of base voltage there will be a voltage atwhich the collector currents of Q₁ and Q₂ are equal. At this voltage thecurrent in Q₁₄ will balance the current in Q₁ and the base of Q₁₂ willbe held at a voltage which maintains the circuit output and the Q₁ - Q₂base voltage constant. Changes in output loading or other disturbanceswhich tend to change the output voltage will change the voltage on thebases of Q₁ and Q₂. This will disturb their collector current balance soas to drive Q₁₂ to restore the output voltage. This control loop forcingthe collector currents of Q₁ and Q₂ to be equal satisfies the condition,previously described, to hold constant C₁ =1.

With the collector currents of Q₁ and Q₂ forced to be equal, the voltagedrop across R22 will be (kT/q) ln J₁ /J₂ =(kT/Q) ln 8. The current inR21 will be just twice that in R22 so that the voltage across R21 willbe proportional to the drop across R22. Therefore, the voltage at thebase of Q₁ which results in the balance condition is the sum of theV_(BE) of Q₁ and the temperature-dependent voltage on R21. This voltageis set (by selecting the ratio of R21 and R22) so that this voltage isjust above the bandgap voltage and satisfies the conditions previouslyoutlined for zero temperature coefficient.

The stabilized base voltage of Q₁ is a fraction of the circuit outputvoltage determined by R31 and R24. The output voltage is, therefore, atemperature stable multiple of the bandgap voltage determined by theresistor ratio. The interbase resistor R23 corrects for the offset anddrift due to base current flow in R31. It also corrects for the basespreading resistance of Q₁, as previously noted.

The voltage divider R28 and R29 is connected across the circuit outputvoltage. It is selected to have a Thevenin equivalent output voltagewhich differs from the circuit output voltage by the bandgap voltage.The equivalent resistance at the divider output is set at twice theresistance of R21. Transistor Q₅ is designed to match Q₁. As a result ofthe equivalent voltage and resistance applied across its base andemitter, its emitter and collector currents will be approximately equalto those of Q₁. This current drives the common base of Q₁₆ and Q₁₇, amatched transistor pair. The matched emitter resistors, R32 and R33,force the emitter currents of Q₁₆ and Q₁₇ to be equal and raise theoutput impedance of Q₁₆. This current mirror "reflects" the collectorcurrent of Q₅ down through Q₃, R27 and Q₁₅. A small fraction of thiscurrent drives the base of Q₄ which in turn drives Q₇ and also suppliesthe current for Q₁₃ and Q₁₄. Since the current in the Q₅, Q₁₇, Q₁₆ andQ₃ path approximates the current in Q₂, it is approximately half thecurrent in Q₁₃ and Q₁₄ combined. This combined current is the majorityof the emitter current in Q₄. By making the emitter area of Q₄ twicethat of Q₃, the current densities and hence the base-emitter voltages ofQ₃ and Q₄ are made nearly equal. Therefore, the voltage at the top ofR27 approximately equals the voltage applied to R25 and R26. Thecurrents in R25, R26 and R27 are approximately equal so that the voltagedrops across them are approximately equal. Similarly, Q₁₅ is sized sothat its emitter current density approximates that of Q₁₃ and Q₁₄. Inthis way the base voltage of Q₁₅ is made nearly equal to the basevoltage of Q₁₃ and Q₁₄. This equality is translated through thebase-emitter voltage of the matched transistors Q₁₁ and Q₁₂ to thecollectors of Q₂ and Q₁ . This keeps the collector voltages of thesetransistors approximately equal at all temperatures and bias conditions.This minimizes problems resulting from different base width modulationin Q₁ and Q₂ which might result from unbalanced collector voltage.

The bias voltage stabilization also keeps the free collector voltage ofQ₁₅ nearly equal to the base voltage. This helps to insure an equalsplit of the current in the forced beta transistor Q₁₅ (beta=1). Thiscurrent split ensures equal emitter currents in Q₁₁ and Q₁₂, therebyminimizing errors due to differences between their base currents.

The circuit as described so far would have a stable "off" state. Theepitaxial layer FET portion of Q₅ eliminates this possibility. The FETprovides a small starting current that turns on the circuit when voltageis applied. Although it diverts some of the current from R28, it hasonly a small effect on the current delivered to Q₁₇. This total currentis determined largely by the voltage drop across the equivalent R28, R29resistance. The slight change in Q₅ V_(BE) which results from thediverted current is a small fraction of the total voltage applied to R28and R29.

The frequency stability of the output control loop is established byC36. This capacitance rolls off the open-loop gain to unity below thefrequency at which excess phase shift in the PNP's might causeinstability.

Output overload protection is provided by Q₆ and R30. The output currentflows through R30 and produces a small voltage drop across it. In theevent of overload, this voltage will rise and drive Q₆ on. As Q₆ comeson it will divert the drive current from the base of Q₄ into the load.As a result, the output current is limited to that necessary to drive Q₆on by way of R30.

The overall circuit consists of a current input amplifier which operatedthe control loop stabilizing the reference voltage. The amplifier inputcircuit, Q₁₃ and Q₁₄, is bootstrapped to the regulated output. Thisbootstrap connection minimizes the effects of power supply voltagevariation on the amplifier which improves the overall supply voltagerejection of the circuit.

Although several preferred embodiments of the invention have beendescribed hereinabove in detail, it is desired to emphasize that suchdetails have been disclosed for the purpose of illustrating the natureof the invention, and should not be considered as necessarily limitingof the invention which can be expressed in many modified forms to meetparticular requirements.

I claim:
 1. A solid-state temperature-compensated voltage supplycomprising:first and second transistors; a resistor connected betweenthe emitter of said first transistor and the emitter of said secondtransistor, circuit means for furnishing supply voltage to said twotransistors to develop current flow therethrough with the currentthrough said first transistor also flowing through said resistor; meansfor sensing the magnitudes of the respective currents flowing throughsaid two transistors; voltage-control means responsive to the currentssensed by said sensing means and operable to adjust the base potentialsof said transistors to maintain the magnitudes of said transistorcurrents at levels which provide a predetermined non-unity ratio ofcurrent densities within the two transistors and thereby cause thecurrent through said resistor to vary positively with respect totemperature of said two transistors; means for developing a firstvoltage proportional to said resistor current and for combining saidfirst voltage with a second voltage which varies negatively with respectto temperature to produce a combined voltage having minimal overallvariation with respect to temperature; and output means coupled to saidlast-named means and including an output terminal providing an outputvoltage proportional to said combined voltage.
 2. Apparatus as in claim1, wherein said voltage-control means comprises a high-gain amplifierserving as a comparator responsive to signals proportional to saidtransistor currents to produce an output signal corresponding to thedifference between said signals proportional to said currents; andmeanscoupling a voltage proportional to said output signal to the bases ofsaid transistors to drive the base potentials to values providing thedesired ratio of current densities in said transistors.
 3. Apparatus asin claim 2, including a voltage-dividing network coupled in the outputof said amplifier and having a network terminal providing a voltagewhich is a predetermined fraction of the amplifier output; andmeanscoupling said network terminal to the bases of said transistors. 4.Apparatus as in claim 2, wherein said sensing means comprises first andsecond load resistors connected in the collector circuits of saidtransistors, respectively.
 5. Apparatus as in claim 1, wherein saidtransistor bases are connected together to provide equal basepotentials.
 6. Apparatus as in claim 1, wherein said transistor basesare coupled together by resistor means to compensate for the effects ofchange in base current.
 7. A solid-state temperature-compensated voltagesupply comprising:first and second transistors; positive and negativevoltage lines; means coupling one of said voltage lines to thetransistor collectors; a resistor connected in a circuit between theemitter of said first transistor and the other of said voltage lines, tocarry the current flowing through said first transistor; meansconnecting the emitter of said second transistor to the end of saidresistor which is remote from said first transistor emitter; means forsensing the magnitudes of the respective currents flowing through saidfirst and second transistors; voltage-control means responsive to saidtransistor currents and operable to adjust the base potentials of saidtransistors to maintain the magnitudes of said transistor currents atlevels which provide a predetermined non-unity ratio of currentdensities within the two transistors and thereby cause the currentthrough said resistor to vary positively with respect to temperature ofsaid two transistors; means for developing a first voltage proportionalto said resistor current and for combining said first voltage with asecond voltage which varies negatively with respect to temperature toproduce a combined voltage having minimal variation with respect totemperture; and output means coupled to said last-named means andincluding an output terminal providing an output voltage proportional tosaid combined voltage.
 8. Apparatus as in claim 7, including first andsecond load resistors in the collector circuits of said transistors,respectively, to develop voltage drops proportional to the transistorcurrents flowing therethrough;an amplifier having its input terminalsconnected to said collector circuits respectively to receive therefromvoltages proportional to the corresponding collector currents; and meansconnecting the output of said amplifier to the bases of said transistorsto apply thereto a voltage proportional to the amplifier output to drivethe base potentials to a value providing a null voltage at the input ofsaid amplifier.
 9. Apparatus as in claim 8, including a voltage-dividernetwork connected to the output of said amplifier; andmeans coupling anintermediate point of said network to said transistor bases to providethereto a control voltage which is a predetermined fraction of theamplifier output.
 10. A solid-state regulated-voltage supplycomprising:first and second transistors; positive and negative supplyvoltage lines; means coupling one of said supply voltage lines to thecollectors of said two transistors; first and second resistors connectedin series between the emitter of said first transistor and the other ofsaid supply voltage lines to carry the current flowing through saidfirst transistor; means connecting the emitter of said second transistorto the junction between said first and second resistors, whereby saidsecond resistor also carries the current flowing through said secondtransistor; means establishing a predetermined relationship between thebase potentials of said two transistors; circuit means for establishingdifferent current densities in said two transistors with the ratio ofcurrent densities being set at a predetermined value to cause thecurrents through said resistors to vary positively with respect totemperature; and output circuit means connected to the base of saidsecond transistor for developing at an output terminal an output voltageproportional to the voltage across said second resistor combinedserially with the V_(BE) voltage of said second transistor.
 11. In avoltage supply of the type comprising means to produce a first voltagehaving a positive temperature coefficient for combination with a secondvoltage having a negative temperature coefficient so as to develop acombined voltage having a substantially reduced overall temperaturecoefficient;the improvement in said means for producing said firstvoltage having a positive temperature coefficient which comprises; firstand second transistors arranged to conduct respective currentstherethrough; means connecting the bases of said two transistorstogether to provide for tracking of the base potentials; sensing meanscoupled to both of said transistors and responsive to said currentspassing therethrough; voltage-control means coupled to said sensingmeans and having an output circuit for producing a control voltageresponsive to the change in the relative levels of said transistorcurrents; means connecting said output circuit to the base of a leastone of said transistors for automatically adjusting the base voltagethereof responsive to said control voltage so as to maintain the ratioof said transistor currents at a value which provides a non-unity ratioof current densities within said transistors; and means connected to theemitters of both of said transistors to produce a voltage proportionalto the difference in base-to-emitter voltage of said two transistors toserve as said first voltage having a positive temperature coefficient.12. A voltage supply as claimed in claim 11, wherein saidvoltage-control means comprises a high-gain amplifier producing saidcontrol voltage at its output.
 13. A voltage supply as claimed in claim12, including means coupling to said one transistor base a voltageproportional to the output voltage of said amplifier.
 14. A voltagesupply as claimed in claim 13, wherein said coupling means comprisesvoltage-dividing means to couple to said transistor base a voltage whichis a pre-set fraction of the amplifier output voltage.
 15. A voltagesupply as claimed in claim 11, wherein the emitters of said twotransistors have substantially different areas.
 16. A voltage supply asclaimed in claim 15, wherein said transistor currents are maintainedequal.
 17. A solid-state temperature-compensated voltage supplycomprising:first and second transistors arranged to conduct respectivecurrents; voltage means to provide base voltage to said transistor basesto produce current densities therein having a non-unity ratio; circuitmeans including resistance means connected to the emitters of said twotransistors to develop a first voltage proportional to the difference inbase-to-emitter voltages of said transistors and to apply said firstvoltage to the emitter of said second transistor; means coupling thebases of said two transistors together to provide for tracking of thebase potentials; an output terminal; and means coupling said outputterminal to the base of said second transistor to provide at said outputterminal an output voltage proportional to said first voltage combinedwith the base-to-emitter voltage of said second transistor.
 18. Avoltage supply as claimed in claim 17, wherein said voltage meanscomprises an amplifier the input of which is coupled to said twotransistors to receive signals therefrom corresponding to the transistorcurrents; andmeans coupling the output of said amplifier to the bases ofsaid two transistors to automatically maintain the base potentials atthe value which produces the required transistor currents to maintainthe transistor current densities at the desired non-unity ratio.
 19. Avoltage supply as claimed in claim 18, wherein said coupling meanscomprises a voltage dividing network arranged to apply to the base ofsaid second transistor a voltage which is a predetermined fraction ofthe amplifier output voltage, whereby said amplifier output serves assaid output terminal developing an output voltage which is greater thanthe base voltage at said second transistor. .Iadd.
 20. A solid-stateband-gap reference circuit comprising:first and second transistors eachhaving a base, emitter and collector; positive and negative supplyleads; first and second sensing resistor means connected between saidpositive supply lead and said collectors respectively; third and fourthseries resistor means connected between the emitter of said firsttransistor and said negative supply lead; means connecting the emitterof said second transistor to a junction between said third and fourthresistor means; means coupling said transistor bases together to providefor tracking of the base voltages thereof; means to apply a controlvoltage to said bases; said resistor means being arranged to set thecurrents through said transistors at levels providing a non-unity ratioof current densities therein; and amplifier means having a pair of inputterminals coupled respectively to said first and second resistor meansto develop an amplifier input voltage responsive to variations in therelative magnitude of the currents flowing through said two transistorsrespectively, whereby the magnitude of the output of said amplifierreflects a comparison between the base voltage of said transistors and apredetermined band-gap reference voltage developed as a compositevoltage consisting of a first voltage proportional to the difference inbase-to-emitter voltages of said two transistors, and a second voltagecorresponding to the base-to-emitter voltage of said second transistor..Iaddend. .Iadd.
 21. A solid-state band-gap reference device comprising:first and second transistors having their bases coupled together; meansfor supplying commonly controllable and tracking base voltages to saidtwo transistors; means coupled to said transistors for developingtherethrough currents of magnitude providing a non-unity ratio ofcurrent densities in said transistors; means for producing a firstvoltage responsive to the difference between the base-to-emittervoltages of said two transistors; means to connect said first voltage tothe emitter of one of said transistors, said first voltage and thebase-to-emitter voltage of said one transistor defining a compositevoltage compensated for temperature; means to sense the changes inrelative magnitudes of current through said two transistors responsiveto changes in said commonly controllable and tracking base voltages ofsaid two transistors; and an amplifier coupled to said sensing means toproduce an output the magnitude of which reflects a comparison of saidchanges in current through said two transistors produced by changes insaid commonly controllable base voltages. .Iaddend.